Techniques for measuring voltages in a circuit

ABSTRACT

A circuit includes a comparator, a programmable current source, and a control circuit. The comparator is operable to compare an internal supply voltage of the circuit to a reference voltage. The programmable current source is operable to supply a first current for the reference voltage. The control circuit is operable to control the first current through the programmable current source based on an output signal of the comparator.

CROSS REFERENCES TO RELATED APPLICATIONS

This patent application is a divisional of U.S. patent application Ser.No. 13/103,832, filed May 9, 2011, which is a divisional of U.S. patentapplication Ser. No. 12/105,262, filed Apr. 17, 2008, now U.S. Pat. No.7,944,248, both of which are incorporated by reference herein in theirentireties.

FIELD OF THE INVENTION

The present invention relates to electronic circuits, and moreparticularly, to techniques for measuring voltages in a circuit.

BACKGROUND OF THE INVENTION

A supply voltage is transmitted to circuit blocks in an integratedcircuit. The supply voltage supplies charge to circuit blocks in theintegrated circuit.

BRIEF SUMMARY OF THE INVENTION

According to some embodiments of the present invention, a circuitincludes a comparator, a resistor divider, a control circuit, amultiplexer, and a programmable gain amplifier. The comparator isoperable to measure an internal voltage of the circuit based on aselected reference voltage. The resistor divider is operable to generatereference voltages. The control circuit is operable to generate a selectsignal based on an output signal of the comparator. The multiplexer isoperable to select one of the reference voltages from the resistordivider as the selected reference voltage based on the select signal.The programmable gain amplifier is configurable to generate acompensation voltage to compensate for an offset voltage of thecomparator. The compensation voltage is provided to an input of thecomparator.

According to other embodiments of the present invention, a circuitincludes a comparator, a programmable current source, and a controlcircuit. The comparator is operable to compare an internal supplyvoltage of the circuit to a reference voltage. The programmable currentsource is operable to supply a first current for the reference voltage.The control circuit is operable to control the first current through theprogrammable current source based on an output signal of the comparator.

Various objects, features, and advantages of the present invention willbecome apparent upon consideration of the following detailed descriptionand the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates an example of how various components can cause asupply voltage to experience a voltage drop before the supply voltage isprovided to circuit blocks in an integrated circuit.

FIG. 2 illustrates an example of a circuit that measures the supplyvoltage at an internal node of an integrated circuit, according to anembodiment of the present invention.

FIG. 3 illustrates an example of a circuit that measures an internalsupply voltage inside an integrated circuit using a comparator and thatcompensates for a voltage offset between the input terminals of thecomparator, according to an embodiment of the present invention.

FIG. 4 illustrates an example of circuitry that can measure the internaltemperature of an integrated circuit and the supply voltage at aninternal node of the integrated circuit, according to another embodimentof the present invention.

FIG. 5 is a graph that illustrates the voltage across a PN junctiondiode as a function of the temperature of the diode.

FIG. 6 illustrates an example of a measuring circuit that measures aninternal supply voltage or a diode voltage using a comparator and thatcompensates for a voltage offset between the input terminals of thecomparator, according to an embodiment of the present invention.

FIG. 7 is a simplified partial block diagram of a field programmablegate array (FPGA) that can include aspects of the present invention.

FIG. 8 shows a block diagram of an exemplary digital system that canembody techniques of the present invention.

DETAILED DESCRIPTION OF THE INVENTION

FIG. 1 illustrates an example of how various components can cause asupply voltage to experience a voltage drop before the supply voltage isprovided to circuit blocks in an integrated circuit. In FIG. 1, circuitboard 101 includes a field programmable gate array (FPGA) integratedcircuit die 103 that is housed in package 102. FPGA die 103 includesnumerous circuit blocks. Only a small fraction of these circuit blocksare shown in FIG. 1. FIG. 1 illustrates examples of the circuit blocksin FPGA die 103 including power on reset (POR) block 111, a timingsensitive block 112, level-shifter circuit 114, voltage-up regulator115, voltage-down regulator 116, and analog circuit blocks 113, 117, and118. The timing sensitive block 112 can include circuitry that operatesat a slower rate in response to a reduced supply voltage VCC.

FPGA die 103 typically receives multiple supply voltages. Differentsupply voltages can, for example, be provided to input/output (I/O)blocks, pre-driver circuits, programmable logic blocks in the core ofthe FPGA, phase-locked loops, delay-locked loops, and other circuitblocks on FPGA die 103. According to some embodiments of the presentinvention, any supply voltage can be measured from an internal node ofan FPGA or application specific integrated circuit (ASIC) die.

Supply voltage VCC is supplied from a supply voltage source to FPGA die103 through board 101 and package 102. The supply voltage VCCexperiences voltage drops Vboard and Vpkg while being transmitted fromnode 120 through wires in board 101 and package 102. Once inside FPGA103, supply voltage VCC experiences additional voltage drops Vwirebondand Vpwr_bus, while being transmitted through wire bonds and a powerbus, respectively, in FPGA die 103 before reaching internal node 121.The internal supply voltage VCC_INT at internal node 121 may have asignificant voltage drop relative to the supply voltage VCC generated bythe supply voltage source at node 120. Alternatively, the supply voltageVCC can be supplied from a voltage regulator inside the integratedcircuit die.

In the example of FIG. 1, supply voltage VCC_INT at node 121 drives PORblock 111, timing sensitive block 112, analog block 113, level-shifterblock 114, voltage-up regulator 115, and voltage-down regulator 116. Thevoltage drop of VCC at node 121 can adversely affect (e.g., slow down)the operation of these circuit blocks and other circuit blocks in FPGAdie 103. The supply voltage source can be programmed to compensate forthe supply voltage drop between nodes 120 and 121 if the internal supplyvoltage VCC_INT at node 121 is known. However, process variationsbetween FPGA integrated circuit dies that have the same FPGAarchitecture can cause the supply voltage VCC_INT at node 121 to varyfrom die to die. Also, temperature variations can cause VCC_INT at node121 to vary within a single FPGA die. Thus, directly measuring theinternal supply voltage VCC_INT at internal node 121 in one FPGA die atone time often does not provide an accurate indication of VCC_INT atnode 121 in other FPGA dies that have the same FPGA architecture orwithin the same FPGA die at a different temperature.

According to some embodiments of the present invention, techniques areprovided for measuring the supply voltage at an internal node of anintegrated circuit so that any voltage drop in the supply voltage can becompensated for by the supply voltage source. Some embodiments of thepresent invention can be used to measure the supply voltage at aninternal node of an integrated circuit in each integrated circuit diethat is manufactured, so that the supply voltage at the internal nodecan be determined accurately regardless of process variations betweenthe integrated circuit dies. Some embodiments of the present inventioncan be used to measure the supply voltage at an internal node of anintegrated circuit at different points in time during the operation ofthe integrated circuit, so that the supply voltage at the internal nodecan be determined accurately regardless of temperature variations of theintegrated circuit.

FIG. 2 illustrates an example of a circuit 200 that measures the supplyvoltage at an internal node inside an integrated circuit, according toan embodiment of the present invention. Measuring circuit 200 in FIG. 2includes voltage comparator 202, finite state machine (FSM) 203,multiplexer 204, resistor group 205, resistor group 206, band gapreference voltage generator 207, and multiplexer 208. Measuring circuit200 is typically fabricated on an integrated circuit, such as aprogrammable integrated circuit or an application specific integratedcircuit (ASIC). Programmable integrated circuits include FPGAs,programmable logic devices (PLDs), and programmable logic arrays.Circuit 200 measures a supply voltage of the integrated circuit thatcircuit 200 is fabricated on.

Comparator 202 receives a selected reference voltage VREF frommultiplexer 204 at its non-inverting (+) input terminal. Comparator 202receives an internal supply voltage VCC_INT from an internal node of theintegrated circuit at its inverting (−) input terminal. Comparator 202compares selected reference voltage VREF to an internal supply voltageVCC_INT of the integrated circuit that contains circuit 200. Comparator202 can be, for example, an operational amplifier that does not consumecurrent at its input terminals.

Multiplexer 208 is programmed to select the internal supply voltageVCC_INT from an internal supply voltage node inside the integratedcircuit containing circuit 200. Multiplexer 208 receives internal supplyvoltages from two or more internal supply voltage nodes in theintegrated circuit. Multiplexer 208 can be programmed to select supplyvoltage VCC_INT1 from a first internal supply voltage node, supplyvoltage VCC_INT2 from a second internal supply voltage node, or anotherinternal supply voltage from a different node. One or more selectsignals SEL_VCCINT are transmitted to the select input terminals ofmultiplexer 208. Multiplexer 208 selects one of the internal supplyvoltages in response to the logic states of the SEL_VCCINT signals. TheSEL_VCCINT select signals can be generated by FSM 203 or by anothercontrol circuit.

The output voltage of comparator 202 is a digital LOCKED signal. Theoutput voltage of comparator 202 is in a logic high state when theselected reference voltage VREF is equal to the internal supply voltageVCC_INT. The output voltage of comparator 202 is in a logic low statewhen the selected reference voltage VREF is not equal to the internalsupply voltage VCC_INT.

Multiplexer 204 selects the reference voltage VREF from a resistordivider formed by resistor groups 205 and 206. The resistor dividerformed by resistor groups 205 and 206 functions as a voltage divider.The voltage between resistor groups 205 and 206 isDesired_VCC_INT=VBG*(R₂₀₆/(R₂₀₆+R₂₀₅)). In the example of FIG. 2,resistor group 205 has 3 resistors R1-R3, and resistor group 206 has 9resistors R4-R12. Resistors R1-R12 are coupled together in series.Resistors R1-R12 can be, for example, polysilicon resistors ortransistors coupled to function as resistors.

The resistor divider formed by resistors R1-R12 generates multiplereference voltages. A reference voltage is generated between each of theresistors R1-R12. For example, a first reference voltage is generatedbetween resistors R1 and R2, a second reference voltage is generatedbetween resistors R2 and R3, a third reference voltage is generatedbetween resistors R3 and R4, a fourth reference voltage is generatedbetween resistors R4 and R5, etc. The reference voltages generated byresistor divider R1-R12 are transmitted to input terminals ofmultiplexer 204. Multiplexer 204 selects one of the reference voltagesat its input terminals from resistors R1-R12 as the selected referencevoltage VREF.

Band gap reference voltage generator 207 generates a constant band gapreference voltage VBG that remains at substantially the same voltageacross an expected range of process, supply voltage, and temperaturevariations of the integrated circuit. The band gap reference voltage VBGfrom band gap reference voltage generator 207 is applied across resistorgroups 205 and 206. Resistor R12 in group 206 receives a ground voltage,and resistor R1 in group 205 receives VBG.

Although the resistances of resistors R1-R12 change in response toprocess and temperature variations of the integrated circuit, the ratiosof resistors R1-R12 remain constant, because process and temperaturevariations typically change the resistances of resistors R1-R12 by thesame amount. Because the band gap reference voltage VBG and the ratiosof resistors R1-R12 remain constant across temperature, process, andsupply voltage variations, the reference voltages generated by resistorsR1-R12 are constant over temperature, process, and supply voltagevariations in the integrated circuit. The reference voltages provided tothe input terminals of multiplexer 204 from resistors R1-R12 remainsubstantially the same over an expected range of temperature, process,and supply voltage variations in the integrated circuit.

The reference voltage generated by the resistor divider betweenresistors R3 and R4 is the desired internal supply voltageDesired_VCC_INT in the embodiment of FIG. 2. For example, if VBG is 1.2volts, and resistors R1-R12 all have the same resistance, thenDesired_VCC_INT equals 0.9 volts. The desired internal supply voltageDesired_VCC_INT is transmitted to input terminal N/2 of multiplexer 204.

To increase the precision of measuring circuit 200 near the desiredinternal supply voltage Desired_VCC_INT, resistors R3 and R4 can eachgenerate multiple reference voltages. Resistor R3 can comprise an (N+1)number of smaller resistors that can generate an N number of referencevoltages V1+, V2+, . . . , VN+, as shown in FIG. 2. A unique referencevoltage is generated between each adjacent pair of two smaller resistorsin R3. Alternatively, resistor R3 can be a single polysilicon resistor,and the N reference voltages can be generated from N equally spacedcontacts along the length of resistor R3.

The reference voltages generated from resistor R3 are incrementalvoltages. Each incremental reference voltage from R3 represents anincrease of a fixed voltage relative to a lower reference voltage. Forexample, resistor R3 can generate 9 reference voltages of 0.91, 0.92,0.93, 0.94, 0.95, 0.96, 0.97, 0.98, and 0.99 volts that representincrements of 10 millivolts (mV). In this example, a reference voltageof 1.00 volts is generated at the node between resistors R2 and R3.Reference voltages V1+, V2+, . . . , VN+ are transmitted from resistorR3 to the (N/2)−1, (N/2)−2, . . . , 0 input terminals of multiplexer204, respectively.

Resistor R4 can comprise an N+1 number of smaller resistors that cangenerate an N number of reference voltages V1−, V2−, . . . , VN−, asshown in FIG. 2. A unique reference voltage is generated between eachadjacent pair of two smaller resistors in R4. Alternatively, resistor R4can be a single polysilicon resistor, and the N reference voltages canbe generated from N equally spaced contacts along the length of resistorR4.

The reference voltages generated from resistor R4 are also incrementalvoltages. Each incremental reference voltage from R4 represents anincrease of a fixed voltage relative to a lower reference voltage. Forexample, resistor R4 can generate 9 reference voltages of 0.81, 0.82,0.83, 0.84, 0.85, 0.86, 0.87, 0.88, and 0.89 volts that representincrements of 10 millivolts (mV). Reference voltages V1−, V2−, . . . ,VN− are transmitted from resistor R4 to the (N/2)+1, (N/2)+2, . . . , Ninput terminals of multiplexer 204, respectively. Multiplexer 204 alsohas additional input terminals for receiving reference voltagesgenerated between resistors R1-R3 and R4-R12.

The N reference voltages generated from resistor R3 and the N referencevoltages generated from resistor R4 also remain at substantially thesame voltages over an expected range of process, supply voltage, andtemperature (PVT) variations of the integrated circuit, because the bandgap reference voltage VBG and the resistance ratios of resistors R1-R12(including the smaller resistors in R3 and R4) remain constant over theexpected range of PVT variations.

Finite state machine (FSM) 203 receives the LOCKED output voltage fromcomparator 202 at a first input terminal of FSM 203. FSM 203 alsoreceives an enable signal IRS_EN at a second input terminal of FSM 203.FSM 203 functions as a control circuit that generates a set of digitalselect signals IR_OUT at its output terminals. The number of IR_OUTselect signals generated by FSM 203 equals log₂ Q. Q equals the numberof input terminals of multiplexer 204 that receive reference voltagesfrom resistor groups 205 and 206, and log refers to a logarithmfunction.

The IR_OUT select signals generated by FSM 203 are transmitted to theselect input terminals of multiplexer 204. Multiplexer 204 selects oneof the Q reference voltages from the resistor divider formed byresistors R1-R12 based on the digital logic states of the IR_OUT selectsignals. Multiplexer 204 transmits the selected reference voltage fromthe resistor divider formed by resistors R1-R12 to the non-invertinginput terminal of comparator 202 as selected reference voltage VREF.

FSM 203 can be, for example, a counter circuit. When FSM 203 senses alogic low state in the IRS_EN signal, FSM 203 may reset the binary valueof its IR_OUT output signals to zero. Thus, each of the output signalsof FSM 203 is in a logic low state when IRS_EN is low.

To begin the operation of measuring circuit 200, the logic state of theIRS_EN enable signal is changed from a logic low state to a logic highstate. After FSM 203 senses a logic high state in the IRS_EN signal, FSM203 beings to increase the binary value of its IR_OUT output signals.FSM 203 increases the binary value of its IR_OUT output signals by 1 ineach subsequent time interval to generate a sequence of increasingbinary values. For example, FSM 203 can increase the IR_OUT outputsignals from 00001, to 00010, to 00011, to 00100, to 00101, to 00111, to01000, to 01001 etc.

Alternatively, FSM 203 can be programmed to initially reset the binaryvalue of the IR_OUT signals to a predefined maximum value (e.g., all 1s)in response to the IRS_EN enable signal being in a logic low state.After the IRS_EN enable signal changes to a logic high state, FSM 203decreases the binary value of its IR_OUT output signals by 1 in eachsubsequent time interval to generate a sequence of decreasing binaryvalues.

Each unique binary value of the IR_OUT select signals causes multiplexer204 to select a unique reference voltage from resistors R1-R12 as theselected reference voltage VREF. FSM 203 generates a sequence of binaryvalues of the IR_OUT select signals that cause multiplexer 204 to selectthe reference voltages generated by resistors R1-R12 as VREF inincreasing order or in decreasing order. For example, an IR_OUT binaryvalue of 00001 may cause multiplexer 204 to select the smallestreference voltage generated by R1-R12 as VREF, an IR_OUT binary value of00010 may cause multiplexer 204 to select the second smallest referencevoltage generated by R1-R12 as VREF, an IR_OUT binary value of 00011 maycause multiplexer 204 to select the third smallest reference voltagegenerated by R1-R12 as VREF, etc. As another example, an IR_OUT binaryvalue of 00001 may cause multiplexer 204 to select the largest referencevoltage generated by R1-R12 as VREF, an IR_OUT binary value of 00010 maycause multiplexer 204 to select the second largest reference voltagegenerated by R1-R12 as VREF, an IR_OUT binary value of 00011 may causemultiplexer 204 to select the third largest reference voltage generatedby R1-R12 as VREF, etc.

Multiplexer 204 causes the selected reference voltage VREF to equalreference voltages generated by resistors R1-R12 in response to theIR_OUT signals. Multiplexer 204 can cause VREF to increase from thesmallest reference voltage to the larger reference voltages generated byresistors R1-R12 in increasing order. Alternatively, multiplexer 204 cancause VREF to decrease from the largest reference voltage to the smallerreference voltages generated by resistors R1-R12 in decreasing order.

The LOCKED output voltage of comparator 202 changes from a logic lowstate to a logic high state after VREF becomes equal to VCC_INT. Afterthe LOCKED output voltage of comparator 202 changes states from a logiclow to a logic high, FSM 203 stops changing the logic states of theIR_OUT signals, and FSM 203 maintains the logic states of the IR_OUTsignals constant. The logic states of the IR_OUT signals remain constantuntil the IRS_EN enable signal changes to a logic low state, whichresets the IR_OUT signals.

When the LOCKED output voltage of comparator 202 is in a logic highstate, the binary value of the IR_OUT signals generated by FSM 203indicates the reference voltage that equals (or approximately equals)the internal supply voltage VCC_INT. The reference voltage selected bymultiplexer 204 from resistors R1-R12 when the LOCKED output voltage ofcomparator 202 is in a logic high state equals (or approximately equals)the internal supply voltage VCC_INT.

The IR_OUT select signals generated by FSM 203 are transmitted toexternal input/output (I/O) pins 211 of the integrated circuit. TheLOCKED output voltage generated by comparator 202 is transmitted toexternal I/O pin 210 of the integrated circuit. The IR_OUT and LOCKEDsignals do not need to be driven to dedicated pins. For example, in anFPGA, the LOCKED and IR_OUT signals can be transmitted to I/O pins210-211 through multiplexers in input/output elements (IOEs) that arealso configurable to transmit signals from programmable logic blocks topins 210-211. Measuring circuit 200 can be used to measure VCC_INTduring the user mode of an FPGA.

A user of the integrated circuit can measure the voltages of the IR_OUTsignals at IR_OUT pins 211 after the LOCKED signal from comparator 202changes states from a logic low to a logic high at pin 210 to determinethe value of the internal supply voltage VCC_INT. As mentioned above,each unique binary value of the IR_OUT select signals causes multiplexer204 to select a unique reference voltage from resistors R1-R12. Afterthe LOCKED output voltage of comparator 202 changes state from a logiclow to a logic high, the logic states of the IR_OUT select signalscorrespond to a reference voltage that equals (or approximately equals)VCC_INT. The precise value of the internal supply voltage VCC_INT may bebetween the final reference voltage VREF selected by multiplexer 204 andthe second to last reference voltage VREF selected by multiplexer 204.

FIG. 3 illustrates an example of a circuit that measures an internalsupply voltage inside an integrated circuit using a comparator and thatcompensates for a voltage offset between the input terminals of thecomparator, according to an embodiment of the present invention.Measuring circuit 300 in FIG. 3 includes comparator 202, finite statemachine 303, multiplexer 204, resistor group 205, resistor group 206, aprogrammable unity gain amplifier 301, and multiplexers 311-314.Measuring circuit 300 is typically fabricated on an integrated circuit,such as an ASIC or programmable integrated circuit.

Finite state machine (FSM) 303 functions as a control circuit thatgenerates a set of output select signals IR_OUT. FSM 303 also generatesfour calibration signals. FSM 303 generates calibration signals OFFSET+,OFFSET−, HOLD, and OFFSET_CAL_DONE. The OFFSET+ calibration signal istransmitted to the select input terminal of multiplexer 312, to thefirst select input terminal of multiplexer 313, and to the firstprogrammable input terminal of amplifier 301. The OFFSET− calibrationsignal is transmitted to the select input terminal of multiplexer 311,to the second select input terminal of multiplexer 313, and to thesecond programmable input terminal of amplifier 301. The HOLDcalibration signal is transmitted to a hold input terminal of amplifier301. The OFFSET_CAL_DONE calibration signal is transmitted to the selectinput terminal of multiplexer 314. An internal supply voltage VCC_INT istransmitted to the 1 input terminal of multiplexer 314, and a desiredinternal supply voltage Desired_VCC_INT is transmitted to the 0 inputterminal of multiplexer 314.

During a calibration mode of measuring circuit 300, programmable unitygain amplifier 301 can be programmed to generate a compensation voltageVCOMP that compensates for an offset voltage in comparator 202 betweenthe input terminals of comparator 202. FSM 303 begins the calibrationmode of measuring circuit 300 by pulling the OFFSET_CAL_DONE signal to alogic low state. When OFFSET_CAL_DONE is in a logic low state,multiplexer 314 selects the desired internal supply voltageDesired_VCC_INT from node 320 between resistor group 205 and resistorgroup 206. Multiplexer 314 transmits Desired_VCC_INT from node 320through its 0 input terminal to the 0 input terminal of multiplexer 311and the 01 input terminal of multiplexer 313.

FSM 303 generates logic states for the IR_OUT signals during thecalibration mode that cause multiplexer 204 to transmit the desiredinternal supply voltage Desired_VCC_INT from node 320 to the 0 inputterminal of multiplexer 312 and the 10 input terminal of multiplexer313. Initially, FSM 303 causes the OFFSET+ and OFFSET− calibrationsignals to be in logic low states. When the OFFSET+ calibration signalis in a logic low state, multiplexer 312 transmits the Desired_VCC_INTvoltage (VREF) from the output terminal of multiplexer 204 to thenon-inverting (+) input terminal of comparator 202. When the OFFSET−calibration signal is in a logic low state, multiplexer 311 transmitsthe Desired_VCC_INT voltage from the output terminal of multiplexer 314to the inverting (−) input terminal of comparator 202. When both OFFSET+and OFFSET− calibration signals are in logic low states, programmablegain amplifier 301 is unused and disabled to save power.

When the voltages at both input terminals of comparator 202 are equal tothe Desired_VCC_INT voltage, the output voltage of comparator 202 isindicative of any offset voltage in comparator 202. For example, ifcomparator 202 has an offset voltage that adds to the voltage at one ofits input terminals relative to the voltage at its other input terminal,then the output voltage of comparator 202 is in a logic low state whenDesired_VCC_INT is applied to both of its input terminals. An offsetvoltage in comparator 202 can be caused by process and layout variationsin the integrated circuit die.

During the calibration mode, FSM 303 sets the logic states of theOFFSET+ and OFFSET− calibration signals in response to the LOCKED outputvoltage of comparator 202 to adjust the gain of programmable unity gainamplifier 301 so that amplifier 301 compensates for any offset voltagein comparator 202. For example, if the output voltage of comparator 202is in a logic low state after the calibration mode begins, FSM 303 candrive the OFFSET+ signal to a logic high state and the OFFSET− signal toa logic low state. While OFFSET+ is high, and OFFSET− is low,multiplexer 313 transmits the Desired_VCC_INT voltage from multiplexer204 through its 10 input terminal to the VIN input terminal of amplifier301 as input voltage VSL.

The gain of amplifier 301 is determined by the logic states of theOFFSET+, OFFSET−, and HOLD signals. When amplifier 301 senses logic lowstates in the HOLD and OFFSET− signals and a logic high state in theOFFSET+ signal, amplifier 301 increases its gain above unity. Amplifier301 amplifies voltage VSL at input terminal VIN to generate acompensation voltage VCOMP at the output terminal of amplifier 301.Voltage VCOMP is larger than the Desired_VCC_INT voltage frommultiplexer 204 by a small amount. Multiplexer 312 transmits theamplified output voltage VCOMP of amplifier 301 to the non-invertinginput terminal of comparator 202 in response to a logic high state inOFFSET+. Multiplexer 311 continues to transmit the Desired_VCC_INTvoltage from multiplexer 314 to the inverting input terminal ofcomparator 202 in response to a logic low state in OFFSET−.

FSM 303 then senses whether the output voltage of comparator 202 changesfrom a logic low state to a logic high state in response to thecompensation voltage VCOMP from amplifier 301. If FSM 303 detects theoutput voltage of comparator 202 changes from a logic low state to alogic high state, then the offset is at the non-inverting input terminalof comparator 202 relative to the inverting input terminal of comparator202. After the output voltage of comparator 202 changes to a logic highstate, FSM 303 causes the HOLD signal to change from a logic low stateto a logic high state to signify to programmable amplifier 301 to haltthe gain increment at that point. Then, the calibration mode iscompleted.

If FSM 303 does not detect the output voltage of comparator 202 changingfrom a logic low state to a logic high state after a predetermined time,the offset is at the inverting input terminal of comparator 202 relativeto the non-inverting input terminal of comparator 202. FSM 303 thenchanges OFFSET+ to a logic low state and OFFSET− to a logic high state.FSM 303 maintains the HOLD signal in a logic low state. In response to alogic low state in OFFSET+, multiplexer 312 transmits voltage VREF tothe non-inverting input terminal of comparator 202. In response to alogic high state in the OFFSET− signal, multiplexer 313 transmits theDesired_VCC_INT voltage from multiplexer 314 to the VIN input terminalof amplifier 301 as voltage VSL, amplifier 301 amplifies voltage VSL togenerate voltage VCOMP, and multiplexer 311 transmits voltage VCOMP tothe inverting input terminal of comparator 202. Programmable amplifier301 increases its gain above unity again in search of the right amountto compensate the offset. Amplifier 301 may continue to increase itsgain as long as one of signals OFFSET+ or OFFSET− is in a logic highstate. If FSM 303 detects the output voltage of comparator 202 changingfrom a logic low state to a logic high state, FSM 303 then changes theHOLD signal from a logic low state to a logic high state to signify toprogrammable amplifier 301 to halt the gain increment at that point.Then, the calibration mode is completed.

In some embodiments, FSM 303 can add compensation to the inverting inputterminal of comparator 202 first. According to some embodiments, FSM 303can lower the gain of programmable amplifier 301 in search of the rightamount of compensation instead of increasing the gain of amplifier 301.

When the calibration mode is completed, FSM 303 pulls theOFFSET_CAL_DONE signal to a logic high state, causing multiplexer 314 totransmit VCC_INT to the 0 input terminal of multiplexer 311 and to the01 input terminal of multiplexer 313. FSM 303 maintains the OFFSET+,OFFSET−, and HOLD signals in logic states that are selected to cancelout or reduce the offset voltage in comparator 202 after the calibrationmode.

FSM 303 resets the binary value of the IR_OUT signals in response to alogic low state in the enable signal IRS_EN. Measuring circuit 300enters a measurement mode after IRS_EN changes to a logic high state.During the measurement mode, FSM 303 increments or decrements the binaryvalue of the IR_OUT select signals, causing multiplexer 204 to selectreference voltages from resistor groups 205 and 206 as VREF inincreasing or decreasing order, as described above with respect to FIG.2.

Using compensation being added to the non-inverting input terminal ofcomparator 202 as an example, FSM 303 keeps the OFFSET+ and HOLD signalshigh and the OFFSET− signal low during the measurement mode. As aresult, multiplexer 313 transmits the output voltage VREF of multiplexer204 to the VIN input terminal of amplifier 301, multiplexer 312transmits the compensation voltage VCOMP from amplifier 301 to thenon-inverting input terminal of comparator 202, and multiplexers 311 and314 transmit VCC_INT to the inverting input terminal of comparator 202.Comparator 202 compares internal supply voltage VCC_INT to each of thereference voltages selected by multiplexer 204 after the referencevoltages have been amplified by amplifier 301. Amplifier 301 reduces orcancels the offset voltage in comparator 202. Signals LOCKED and IR_OUTare transmitted to pins 210-211.

Using compensation being added to the inverting input terminal ofcomparator 202 as another example, FSM 303 keeps the OFFSET+ signal in alogic low state and drives the OFFSET− and HOLD signals to logic highstates during measurement mode. As a result, multiplexer 313 transmitsthe output voltage VCC_INT of multiplexer 314 to the VIN input terminalof amplifier 301, multiplexer 312 transmits VREF to the non-invertinginput terminal of comparator 202, and multiplexer 311 transmits VCOMPfrom amplifier 301 to the inverting input terminal of comparator 202.During the measurement mode, amplifier 301 amplifies VCC_INT to generatethe compensation voltage VCOMP. Comparator 202 compares each of thereference voltages selected by multiplexer 204 to the voltage of VCC_INTamplified by amplifier 301. Amplifier 301 reduces or cancels the offsetvoltage in comparator 202.

Measuring circuit 300 typically calibrates amplifier 301 to compensatefor an voltage offset in comparator 202 only one time before VCC_INT ismeasured. Measuring circuit 300 does not need to perform the calibrationmode prior to each measurement of VCC_INT to recalibrate amplifier 301.

FIG. 4 illustrates an example of circuitry that can measure the internaltemperature of an integrated circuit and the supply voltage at aninternal node of the integrated circuit, according to another embodimentof the present invention. Measuring circuit 400 includes an 8-bit finitestate machine (FSM) 401, a programmable current source 402, a voltagecomparator 403, a multiplexer 404, a current source 405, a diode 406, aresistor 407, and I/O pin 408. One terminal of resistor 407 is coupledto the inverting input terminal (−) of comparator 403, and the otherterminal of resistor 407 receives the ground voltage. Measuring circuit400 is typically fabricated on an integrated circuit, such as an ASIC ora programmable integrated circuit.

The voltage between programmable current source 402 and resistor 407 isa reference voltage VREF. The reference voltage VREF is transmitted tothe inverting (−) input terminal of comparator 403. 8-bit FSM 401generates 8 TSD_IRS[7:0] control signals. The logic states of the 8digital TSD_IRS[7:0] control signals determine the amount of currentgenerated by programmable current source 402. The amount of currentgenerated by programmable current source 402 affects the referencevoltage VREF. FSM 401 functions as a control circuit.

Multiplexer 404 receives select signal TSD_IRS_SEL at its select inputterminal. When TSD_IRS_SEL is in a logic high state, multiplexer 404transmits an internal supply voltage VCC_INT through its 1 inputterminal to the non-inverting input terminal (+) of comparator 403 asvoltage VSEL. When TSD_IRS_SEL is in a logic low state, multiplexer 404transmits the voltage across diode 406 through its 0 input terminal tothe non-inverting input terminal of comparator 403 as voltage VSEL.Current source 405 drives a current through diode 406. The voltageacross diode 406 at the 0 input terminal of multiplexer 404 isreferenced to the ground voltage.

In one embodiment, FSM 401 decreases the binary value of theTSD_IRS[7:0] control signals, causing the current through programmablecurrent source 402 to increase until the reference voltage VREF equalsVSEL. In another embodiment, FSM 401 increases the binary value of theTSD_IRS[7:0] control signals, causing the current through programmablecurrent source 402 to decrease until the reference voltage VREF equalsVSEL.

Comparator 403 compares voltage VSEL to reference voltage VREF togenerate a LOCKED output voltage that is transmitted to I/O pin 408 andFSM 401. The output voltage of comparator 403 is in a logic high statewhen voltage VSEL equals reference voltage VREF. The output voltage ofcomparator 403 is in a logic low state when voltage VSEL does not equalreference voltage VREF.

FSM 401 receives the LOCKED output voltage of comparator 403 at an inputterminal. When the output voltage of comparator 403 changes from a logiclow state to a logic high state, FSM 403 maintains the logic states ofthe TSD_IRS[7:0] control signals constant. As a result, the final logicstates of the TSD_IRS[7:0] control signals are indicative of voltageVSEL, which equals either internal supply voltage VCC_INT or the voltageacross diode 406, depending on the logic state of the TSD_IRS_SELsignal.

The voltage across a PN junction diode varies as a function of itstemperature, as shown in FIG. 5. Each temperature produces a uniquevoltage across the diode. The voltage across a PN junction diodetypically varies linearly across the temperature range shown in FIG. 5.Diode 406 in measuring circuit 400 functions as a temperature sensingdiode (TSD). Measuring circuit 400 can compare the voltage across diode406 to reference voltage VREF, as described above, to generate logicstates of the TSD_IRS[7:0] signals that are indicative of thetemperature of diode 406 and the temperature of the integrated circuitcontaining circuit 400. The temperature of a diode is typically anaccurate measure of the temperature of the integrated circuit thatcontains the diode.

FIG. 6 illustrates an example of a measuring circuit 600 that measuresan internal supply voltage or a diode voltage using a comparator andthat compensates for a voltage offset between the input terminals of thecomparator, according to an embodiment of the present invention.Measuring circuit 600 as shown in FIG. 6 includes 8-bit finite statemachine (FSM) 401, voltage comparator 403, multiplexer 404, diode 406,resistor 407, multiplexers 601-608, inverter 609, offset calibrationfinite state machine (FSM) 610, offset adjust logic 611, p-channel metaloxide semiconductor field-effect transistors (MOSFETs) 616-619,resistors 620-623, band gap reference voltage divider 612, andinput/output (I/O) pins 408 and 614. Measuring circuit 600 is typicallyfabricated on an integrated circuit, such as an ASIC or a programmableintegrated circuit.

Programmable current source 402 includes eight p-channel MOSFETs 616 andeight resistors 620. P-channel MOSFETs 616 are binary weighted. Forexample, transistors 616 can have relative sizes of 1x, 2X, 4X, 8X, 16X,32X, 64X, and 128X, where X represents a transistor channelwidth-to-length ratio. Resistors 620 are also binary weighted, e.g., 1Y,2Y, 4Y, 8Y, 16Y, 32Y, 64Y, 128Y, where Y represents a resistance value.

Offset current source 624 includes five p-channel MOSFETs 619 and fiveresistors 623. P-channel transistors 619 are binary weighted. Forexample, transistors 619 can have relative sizes of 1X, 2X, 4X, 8X, and16X. Resistors 623 are also binary weighted (e.g., 1Y, 2Y, 4Y, 8Y, 16Y).Each of p-channel transistors 616-619 receives supply voltage VCC at asource terminal and is coupled to one of the resistors. Resistors620-623 are coupled to the inverting input terminal (−) of comparator403 and to the p-channel transistors, as shown in FIG. 6.

Measuring circuit 600 can measure the supply voltage VCC_INT at internalnode 613 of the integrated circuit that contains circuit 600. Measuringcircuit 600 can also measure the voltage across a temperature sensingdiode (TSD) 406. Techniques for accurately measuring the voltage acrossdiode 406 or the internal supply voltage VCC_INT are now described.

Process and layout variations in the integrated circuit may causecomparator 403 to have an offset voltage between its inverting (−) andnon-inverting (+) input terminals. Measuring circuit 600 containscalibration circuitry that can compensate for an offset voltage betweenthe input terminals of comparator 403 during a calibration mode. Thecalibration circuitry includes offset calibration FSM 610, offset adjustlogic 611, offset current source 624, multiplexers 601-608, transistor618, and resistor 622.

Offset calibration FSM 610 begins a calibration mode of measuringcircuit 600 by pulling the OFFSETCALDONE signal to a logic low state. Alogic low state in the OFFSETCALDONE signal causes multiplexers 603-605to select the voltages at their 0 input terminals. During thecalibration mode, offset calibration FSM 610 adjusts the voltage VREF atthe inverting (−) input terminal of comparator 403 to compensate for anoffset voltage between the input terminals of comparator 403.

Band gap reference voltage divider circuit 612 generates three referencevoltages. The three reference voltages are VBGP7558V, VBGP6572V, andDesired_VCC_INT. Band gap reference voltage divider circuit 612generates these three reference voltages using a resistor dividercircuit that receives a band gap voltage VBG from a band gap referencevoltage generator. The band gap voltage VBG generated by the band gapreference voltage generator remains substantially constant over anexpected range of process, supply voltage, and temperature variations ofthe integrated circuit. Also, the ratios of the resistors in theresistor divider in band gap reference voltage divider circuit 612remain substantially constant over an expected range of process, supplyvoltage, and temperature variations of the integrated circuit. As aresult, reference voltages VBGP7558V, VBGP6572V, and Desired_VCC_INTremain substantially constant over the expected range of process, supplyvoltage, and temperature variations of the integrated circuit.

During the measurement mode, measuring circuit 600 can be used tomeasure the voltage across diode 406 or the supply voltage VCC_INT atinternal node 613 of the integrated circuit that contains circuit 600. Adigital signal TSD_IRS_SEL controls whether measuring circuit 600measures the voltage across diode 406 or the supply voltage VCC_INT atinternal node 613 of the integrated circuit. The TSD_IRS_SEL signal istransmitted to the select input terminals of multiplexers 602, 606, and404. The TSD_IRS_SEL signal is also transmitted to an input terminal ofoffset calibration FSM 610.

The TSD_IRS_SEL signal is pulled to a logic low state before ameasurement of the voltage across diode 406. A logic low state in theTSD_IRS_SEL signal causes multiplexers 602, 606, and 404 to select thevoltages at their 0 input terminals. The TSD_IRS_SEL signal is alsotransmitted to inverter 609. When the TSD_IRS_SEL signal is in a logiclow state, the output voltage of inverter 609 is in a logic high state,and p-channel transistor 617 is off.

Measuring circuit 600 is calibrated during a calibration mode after theTSD_IRS_SEL signal is pulled to a logic low state, but before thevoltage across diode 406 is measured. Measuring circuit 600 can becalibrated at an expected operating temperature of the integratedcircuit that contains circuit 600 during the calibration mode. FIG. 6,for example, contains circuitry that can be used to calibrate circuit600 at an expected temperature of 25° C. or 85° C.

If a user decides to calibrate circuit 600 using the voltage acrossdiode 406 at 25° C., then a select signal RREF25C85C is pulled to alogic low state. Select signal RREF25C85C controls multiplexers 601 and607. When the RREF25C85C, TSD_IRS_SEL, and OFFSETCALDONE signals are allin logic low states during the calibration mode, multiplexers 605-607transmit reference voltage VBGP7558V to the non-inverting (+) inputterminal of comparator 403. Reference voltage VBGP7558V has a constantvoltage of 0.7558 volts, which is the voltage across diode 406 at atemperature of 25° C. Also, when the RREF25C85C, TSD_IRS_SEL, andOFFSETCALDONE signals are all in logic low states during the calibrationmode, multiplexers 601-603 transmit 8-bit reference code RREF25C[7:0](e.g., 1001 1001) to the gates of transistors 616 in programmablecurrent source 402.

If a user decides to calibrate circuit 600 using the voltage acrossdiode 406 at 85° C., then the select signal RREF25C85C is pulled to alogic high state. When the RREF25C85C signal is in a logic high state,and the TSD_IRS_SEL and OFFSETCALDONE signals are in logic low statesduring the calibration mode, multiplexers 605-607 transmit referencevoltage VBGP6572V to the non-inverting (+) input terminal of comparator403. Reference voltage VBGP6572V has a constant voltage of 0.6572 volts,which is the voltage across diode 406 at a temperature of 85° C. Also,when the RREF25C85C signal is in a logic high state, and the TSD_IRS_SELand OFFSETCALDONE signals are in logic low states during the calibrationmode, multiplexers 601-603 transmit 8-bit reference code RREF85C[7:0](e.g., 1101 0101) to the gates of transistors 616 in programmablecurrent source 402.

Either reference voltage VBGP7558V or VBGP6572V is transmitted to thenon-inverting input terminal of comparator 403 during a calibration modethat precedes a measurement of the voltage across diode 406.

Each of the 8-bit reference codes RREF25C[7:0] and RREF85C[7:0] have 8digital signals that are transmitted in parallel through a bus. Each ofthe multiplexers 601-603 represents eight 2-to-1 multiplexers that areconfigurable to select the 8 digital signals in one of the 8-bitreference codes.

Measuring circuit 600 can also be used to measure the supply voltageVCC_INT at internal node 613 of the integrated circuit that containscircuit 600. To begin a measurement of internal supply voltage VCC_INT,the TSD_IRS_SEL signal is pulled to a logic high state. When theTSD_IRS_SEL signal is in a logic high state, multiplexers 602, 606, and404 select the voltages at their 1 input terminals, and the outputvoltage of inverter 609 is in a logic low state, which causes p-channeltransistor 617 to be on.

Measuring circuit 600 is calibrated during a calibration mode after theTSD_IRS_SEL signal is pulled to a logic high state, but before internalsupply voltage VCC_INT is measured. When TSD_IRS_SEL is in a logic highstate and OFFSETCALDONE is in a logic low state during the calibrationmode, multiplexers 605-606 transmit the reference voltageDesired_VCC_INT from circuit 612 to the non-inverting input terminal ofcomparator 403, and multiplexers 602-603 transmit a predefined 8-bitreference code (e.g., 0100 0000) to the gates of transistors 616 inprogrammable current source 402.

The 8-bit reference code selected by multiplexer 603 is transmitted tothe gates of the eight p-channel transistors 616 in 8-bit current source402. Each of the 8 digital signals in the 8-bit reference code controlsthe gate voltage of one of transistors 616. The 8-bit reference codeselected by multiplexer 603 turns on one or more of transistors 616 tosupply current through resistors 620 and 407. The signals in the 8-bitreference code that are in logic low states turn on the transistors 616that they control, and the signals in the 8-bit reference code that arein logic high states turn off the transistors 616 that they control.

During the calibration mode, offset calibration FSM 610 turns on one ormore of p-channel transistors 618 and 619 to supply current through oneor more of resistors 622 and 623, respectively. FSM 610 pulls signalCALP to a logic low state to turn on transistor 618. FSM 610 pullssignal CALP to a logic high state to turn off transistor 618. FSM 610also generates a 5-bit calibration code OFFSETSM[4:0]. 5-bit calibrationcode OFFSETSM[4:0] is transmitted to the gates of the five p-channeltransistors 619 through multiplexer 604 when the OFFSETCALDONE signal isin a logic low state. Thus, OFFSETSM[4:0] controls the conductive statesof transistors 619 during the calibration mode. Each bit inOFFSETSM[4:0] that is in a logic high state turns off a transistor 619,and each bit in OFFSETSM[4:0] that is in a logic low state turns on atransistor 619.

Resistors 620-623 and resistor 407 create a variable resistor dividerbetween supply voltage VCC and ground that functions as a variablevoltage divider. The variable resistor divider generates a referencevoltage VREF at the inverting input terminal of comparator 403. Duringthe calibration mode, the logic states of the 8-bit reference codeselected by multiplexer 603, the CALP signal, the TSD_IRS_SEL signal,and the OFFSETSM[4:0] signals determine the current through transistors616-619 and resistors 407 and 620-623. The current through transistors616-619 and resistors 407 and 620-623 determines the reference voltageVREF at the inverting input terminal of comparator 403.

When TSD_IRS_SEL and RREF25C85C are both in logic low states, the 8-bitreference code RREF25C[7:0], the CALP signal, the TSD_IRS_SEL signal,the OFFSETSM[4:0] signals, and the VBGP7558V voltage signal generate avoltage of 0.7558 volts at both input terminals of comparator 403 in thecalibration mode prior to measuring the voltage across diode 406. WhenTSD_IRS_SEL is in a logic low state and RREF25C85C is in a logic highstate, the 8-bit reference code RREF85C[7:0], the CALP signal, theTSD_IRS_SEL signal, the OFFSETSM[4:0] signals, and the VBGP6572V voltagesignal generate a voltage of 0.6572 volts at both input terminals ofcomparator 403 in the calibration mode prior to a measurement of thevoltage across diode 406.

When TSD_IRS_SEL is in a logic high state, the 8-bit reference code 01000000, the CALP signal, the TSD_IRS_SEL signal, the OFFSETSM[4:0]signals, and the Desired_VCC_INT voltage signal generate a voltage equalto the Desired_VCC_INT voltage (e.g., 0.9 volts) at both input terminalsof comparator 403 during the calibration mode prior to a measurement ofVCC_INT. Thus, the voltages transmitted to both input terminals ofcomparator 403 during the calibration mode are the same voltage.

Transistor 617 and resistor 621 are added to circuit 600 to increase thereference voltage VREF at the inverting input terminal of comparator 403during the supply voltage calibration mode and the supply voltagemeasurement mode.

The offset calibration FSM 610 adjusts the conductive states oftransistors 619 using the 5-bit digital offset code OFFSETSM[4:0] duringthe calibration mode to compensate for an offset voltage between theinput terminals of comparator 403. The output voltage of comparator 403is indicative of any offset voltage in comparator 403 during thecalibration mode. The 5-bit digital offset code OFFSETSM[4:0] output byoffset calibration FSM 610 generates one of 32 different currentsettings for offset current source 624. A change in the binary value of5-bit digital offset code OFFSETSM[4:0] either increases or decreasesvoltage VREF at the inverting input terminal of comparator 403 bychanging the current setting for offset current source 624.

FSM 610 senses the LOCKED output voltage of comparator 403 during thecalibration mode. If the reference voltage VREF at the inverting inputterminal of comparator 403 is initially not equal to the band gapreference voltage from circuit 612 at the non-inverting input terminalof comparator 403, the LOCKED output voltage of comparator 403 is in alogic low state, and FSM 610 changes the binary value of theOFFSETSM[4:0] digital code to equalize reference voltage VREF with theband gap reference voltage from circuit 612. The LOCKED output voltageof comparator 403 is in a logic high state when the voltages at itsinput terminals are equal.

After the LOCKED output voltage of comparator 403 toggles from a logiclow state to a logic high state during the calibration mode, FSM 610ends the calibration mode and begins the measurement mode by pulling theOFFSETCALDONE signal to a logic high state. After the calibration mode,the logic states of the OFFSETSM[4:0] signals generate an offset currentin current source 624 that compensates for any offset voltage betweenthe input terminals of comparator 403. During the measurement mode, FSM610 maintains the OFFSETSM[4:0] signals in constant logic states.

When the OFFSETCALDONE signal is in a logic high state during themeasurement mode, multiplexer 605 transmits the output signal ofmultiplexer 404 to the non-inverting input terminal of comparator 403.If the TSD_IRS_SEL signal is in a logic low state, then multiplexer 404transmits the voltage across diode 406 to the non-inverting inputterminal of comparator 403. If the TSD_IRS_SEL signal is in a logic highstate, then multiplexer 404 transmits internal supply voltage VCC_INTfrom internal node 613 to the non-inverting input terminal of comparator403.

Also, multiplexer 604 transmits the 5-bit output code of offset adjustlogic 611 to the gates of transistors 619 when the OFFSETCALDONE signalis in a logic high state. When OFFSETCALDONE is high, offset adjustlogic 611 transmits either the OFFSETSM[4:0] code or an adjusted code tothe gates of transistors 619. Offset adjust logic 611 generates theadjusted code by adding to or subtracting from the binary value of theOFFSETSM[4:0] code. Offset adjust logic 611 receives a 6-bit codeOFFSET[5:0] from multiplexer 608. One bit OFFSET[5] of the 6-bit codeOFFSET[5:0] indicates whether to add or subtract from the binary valueof OFFSETSM[4:0]. The remaining 4 bits OFFSET[4:0] of the OFFSET[5:0]code indicate the binary value to add to or subtract from the binaryvalue of OFFSETSM[4:0]. Multiplexer 608 selects either the ROFFSET[5:0]code or the OFFSETUSR[5:0] code as the OFFSET[5:0] code in response toselect signal ROFFSETSEL. Thus, the logic signals in the OFFSETSM[4:0]code (or a user adjusted code) control the conductive states oftransistors 619 during the measurement mode based on the binary value ofthe OFFSETSM[4:0] code that was selected by FSM 610 during thecalibration mode.

In addition, when the OFFSETCALDONE signal is in a logic high state,multiplexer 603 transmits the 8-bit control code TSD_IRS[7:0] to thegates of transistors 616 during the measurement mode. Control codeTSD_IRS[7:0] then controls the conductive states of transistors 616 inprogrammable current source 402. 8-bit FSM 401 is enabled when theOFFSETCALDONE signal transitions to a logic high state. When FSM 401 isenabled, FSM 401 resets the binary value of the control codeTSD_IRS[7:0] to a default binary value (e.g., 00000000 or 11111111). FSM401 then decreases (or increases) the binary value of control codeTSD_IRS[7:0] in each subsequent time interval to increase (or decrease)the current through programmable current source 402, until FSM 401senses a change in the LOCKED output voltage of comparator 403 from alogic low state to a logic high state. Comparator 403 and FSM 401together form an analog-to-digital converter circuit.

The LOCKED output voltage of comparator 403 is in a logic high statewhen reference voltage VREF at its inverting input terminal is equal tothe voltage at its non-inverting input terminal (i.e., the voltageacross diode 406 or VCC_INT during measurement mode). After the LOCKEDoutput voltage of comparator 403 changes from a logic low state to alogic high state, FSM 401 maintains the signals in control codeTSD_IRS[7:0] in constant logic states. The logic states of the 8-bitcontrol code TSD_IRS[7:0] generated by FSM 401 after the LOCKED outputvoltage changes from a logic low state to a logic high state areindicative of either the voltage across diode 406 or the internal supplyvoltage VCC_INT, depending on the logic state of TSD_IRS_SEL. Forexample, if TSD_IRS_SEL is a logic high and the binary value of controlcode TSD_IRS[7:0] is greater than 01000000, then VCC_INT is greater thanthe Desired_VCC_INT voltage. As another example, if TSD_IRS_SEL is alogic high and the binary value of control code TSD_IRS[7:0] is lessthan 01000000, then VCC_INT is less than the Desired_VCC_INT voltage.

The control code TSD_IRS[7:0] is transmitted to 8 output pins 614 of theintegrated circuit for analysis. Output pins 614 are driven by circuitryin the input/output elements (IOEs). Output pins 614 do not need to bededicated pins. Output pins 614 can be pins that are shared byprogrammable logic blocks in an FPGA. For example, multiplexers can beconfigured to transmit either control code TSD_IRS[7:0] or digitalsignals from programmable logic blocks to output pins 614.

The 8-bit code TSD_IRS[7:0] generated by FSM 401 allows for 256 possiblecombinations for the conductive states of transistors 616. Thus, 8-bitcontrol code TSD_IRS[7:0] can generate 256 different voltages for VREF.FSM 401 can generate a very finely tuned reference voltage VREF in orderto determine the voltage at the non-inverting input terminal ofcomparator 403 within a high degree of precision. For example, thevoltage across diode 406 varies from 0.570 to 0.870 volts across thetemperature range shown in FIG. 5. This voltage range equals a change of300 mV. Therefore, 256 different voltages for VREF corresponds to avoltage change of about 1 mV in VREF in response to each increase (ordecrease) of 1 in the binary value of 8-bit control code TSD_IRS[7:0].

Programmable current source 402, offset current source 624, and thecurrent sources formed by transistors 617-618 and resistors 621-622 arenon-ideal current sources. Therefore, the currents generated by thesecurrent sources change in response to variations in the process, thesupply voltage, and the temperature of circuit 600. If the temperatureof circuit 600 changes during the operation of circuit 600, then thecurrent through transistors 616-619 may change in response to thetemperature change. Because the temperature of measuring circuit 600 mayhave changed since the last calibration mode, measuring circuit 600 isrecalibrated before each measurement of the voltage across diode 406 orthe internal supply voltage VCC_INT. Measuring circuit 600 isrecalibrated by performing the calibration mode again. FSM 610 beginsthe calibration mode by pulling the OFFSETCALDONE signal to a logic lowstate. FSM 610 then readjusts the logic states of the signals thatcontrol the gate voltages of transistors 618-619 during the calibrationmode to compensate for the offset voltage between the input terminals ofcomparator 403, as described above.

FIG. 7 is a simplified partial block diagram of a field programmablegate array (FPGA) 700 that can include aspects of the present invention.FPGA 700 is merely one example of an integrated circuit that can includefeatures of the present invention. It should be understood thatembodiments of the present invention can be used in numerous types ofintegrated circuits such as field programmable gate arrays (FPGAs),programmable logic devices (PLDs), complex programmable logic devices(CPLDs), programmable logic arrays (PLAs), and application specificintegrated circuits (ASICs).

FPGA 700 includes a two-dimensional array of programmable logic arrayblocks (or LABs) 702 that are interconnected by a network of column androw interconnect conductors of varying length and speed. LABs 702include multiple (e.g., 10) logic elements (or LEs).

An LE is a programmable logic block that provides for efficientimplementation of user defined logic functions. A FPGA has numerouslogic elements that can be configured to implement various combinatorialand sequential functions. The logic elements have access to aprogrammable interconnect structure. The programmable interconnectstructure can be programmed to interconnect the logic elements in almostany desired configuration.

FPGA 700 also includes a distributed memory structure including randomaccess memory (RAM) blocks of varying sizes provided throughout thearray. The RAM blocks include, for example, blocks 704, blocks 706, andblock 708. These memory blocks can also include shift registers and FIFObuffers.

FPGA 700 further includes digital signal processing (DSP) blocks 710that can implement, for example, multipliers with add or subtractfeatures. Input/output elements (IOEs) 712 located, in this example,around the periphery of the chip support numerous single-ended anddifferential input/output standards. Each IOE 712 is coupled to anexternal terminal (i.e., a pin) of FPGA 700. For example, a subset ofthe IOEs 712 can be coupled to pins 408 and 614. It is to be understoodthat FPGA 700 is described herein for illustrative purposes only andthat the present invention can be implemented in many different types ofPLDs, FPGAs, and ASICs.

The present invention can also be implemented in a system that has anFPGA as one of several components. FIG. 8 shows a block diagram of anexemplary digital system 800 that can embody techniques of the presentinvention. System 800 can be a programmed digital computer system,digital signal processing system, specialized digital switching network,or other processing system. Moreover, such systems can be designed for awide variety of applications such as telecommunications systems,automotive systems, control systems, consumer electronics, personalcomputers, Internet communications and networking, and others. Further,system 800 can be provided on a single board, on multiple boards, orwithin multiple enclosures.

System 800 includes a processing unit 802, a memory unit 804, and aninput/output (I/O) unit 806 interconnected together by one or morebuses. According to this exemplary embodiment, FPGA 808 is embedded inprocessing unit 802. FPGA 808 can serve many different purposes withinthe system in FIG. 8. FPGA 808 can, for example, be a logical buildingblock of processing unit 802, supporting its internal and externaloperations. FPGA 808 is programmed to implement the logical functionsnecessary to carry on its particular role in system operation. FPGA 808can be specially coupled to memory 804 through connection 810 and to I/Ounit 806 through connection 812.

Processing unit 802 can direct data to an appropriate system componentfor processing or storage, execute a program stored in memory 804,receive and transmit data via I/O unit 806, or other similar function.Processing unit 802 can be a central processing unit (CPU),microprocessor, floating point coprocessor, graphics coprocessor,hardware controller, microcontroller, field programmable gate arrayprogrammed for use as a controller, network controller, or any type ofprocessor or controller. Furthermore, in many embodiments, there isoften no need for a CPU.

For example, instead of a CPU, one or more FPGAs 808 can control thelogical operations of the system. As another example, FPGA 808 acts as areconfigurable processor that can be reprogrammed as needed to handle aparticular computing task. Alternately, FPGA 808 can itself include anembedded microprocessor. Memory unit 804 can be a random access memory(RAM), read only memory (ROM), fixed or flexible disk media, flashmemory, tape, or any other storage means, or any combination of thesestorage means.

The foregoing description of the exemplary embodiments of the presentinvention has been presented for the purposes of illustration anddescription. The foregoing description is not intended to be exhaustiveor to limit the present invention to the examples disclosed herein. Insome instances, features of the present invention can be employedwithout a corresponding use of other features as set forth. Manymodifications, substitutions, and variations are possible in light ofthe above teachings, without departing from the scope of the presentinvention.

What is claimed is:
 1. A circuit comprising: a comparator to compare aninternal supply voltage of the circuit to a reference voltage; aprogrammable current source to supply a first current for the referencevoltage; a first control circuit to control the first current throughthe programmable current source based on an output signal of thecomparator; an offset current source to supply a second current for thereference voltage; and a second control circuit to control the secondcurrent through the offset current source, wherein the comparatorcompares a desired internal supply voltage to the reference voltageduring a supply voltage calibration mode, and wherein the second controlcircuit adjusts the second current through the offset current sourceduring the supply voltage calibration mode in response to the outputsignal of the comparator to compensate for an offset voltage in thecomparator.
 2. The circuit of claim 1, wherein the circuit is in anintegrated circuit, and wherein the internal supply voltage is providedto the comparator from an internal node of the integrated circuit. 3.The circuit of claim 1 further comprising: a diode; and a firstmultiplexer configurable to provide a voltage across the diode to thecomparator during a temperature measurement mode in response to a selectsignal, wherein the comparator compares the voltage across the diode tothe reference voltage during the temperature measurement mode, andwherein the first multiplexer is configurable to provide the internalsupply voltage to the comparator during a supply voltage measurementmode in response to the select signal.
 4. The circuit of claim 3 furthercomprising: a transistor to supply a third current for the referencevoltage during the supply voltage measurement mode in response to theselect signal, wherein the transistor is off during the temperaturemeasurement mode.
 5. The circuit of claim 3 further comprising: a secondmultiplexer to receive diode reference control signals at first inputs,supply voltage reference control signals at second inputs, and theselect signal at a select input; and a third multiplexer to receivefirst control signals from the first control circuit at first inputs andsecond control signals from the second multiplexer at second inputs,wherein the third multiplexer is configurable to provide a selected setof control signals to gate terminals of transistors in the programmablecurrent source in response to a calibration mode signal.
 6. The circuitof claim 5 further comprising: a fourth multiplexer configurable toprovide a band gap reference voltage to the comparator during a diodevoltage calibration mode, wherein the fourth multiplexer is configurableto provide the desired internal supply voltage to the comparator duringthe supply voltage calibration mode.
 7. The circuit of claim 6 furthercomprising: a fifth multiplexer configurable to provide one of a signalselected by the fourth multiplexer or a signal selected by the firstmultiplexer to the comparator.
 8. An integrated circuit comprising: acomparator to compare a voltage in the integrated circuit to a referencevoltage; a programmable current source to supply a first current for thereference voltage; a first control circuit to control the first currentthrough the programmable current source based on an output signal of thecomparator; a first multiplexer to select a supply voltage or a voltagefrom a diode as a selected signal; and a second multiplexer to providethe selected signal to the comparator as the voltage in the integratedcircuit.
 9. A method for measuring an internal supply voltage, themethod comprising: comparing the internal supply voltage to a referencevoltage to generate a comparison signal; generating a programmablecurrent supplied to the reference voltage; generating first controlsignals for controlling the programmable current during a supply voltagemeasurement mode based on the comparison signal; generating an offsetcurrent supplied to the reference voltage; generating second controlsignals for controlling the offset current; comparing a desired internalsupply voltage to the reference voltage to generate the comparisonsignal during a supply voltage calibration mode; and adjusting thesecond control signals during the supply voltage calibration mode inresponse to the comparison signal to compensate for an offset voltageused to generate the comparison signal.
 10. The method of claim 9,wherein the method is performed by an integrated circuit, and whereinthe internal supply voltage is provided from an internal node of theintegrated circuit.
 11. The method of claim 9 further comprising:generating a diode voltage drop during a temperature measurement mode;and comparing the diode voltage drop to the reference voltage during thetemperature measurement mode, wherein generating first control signalsfor controlling the programmable current during a supply voltagemeasurement mode based on the comparison signal comprises adjusting thefirst control signals until the comparison signal changes state.
 12. Themethod of claim 11 further comprising: supplying a third current for thereference voltage during the supply voltage measurement mode; andturning off the third current during the temperature measurement mode.13. The method of claim 11 further comprising: selecting diode referencecontrol signals or supply voltage reference control signals as thirdcontrol signals using a first multiplexer; and selecting the firstcontrol signals or the third control signals using a second multiplexerto provide a selected set of control signals to gate terminals oftransistors in a programmable current source that generates theprogrammable current.
 14. The method of claim 11 further comprising:using a band gap reference voltage to generate the comparison signalduring a diode voltage calibration mode; and using the desired internalsupply voltage to generate the comparison signal during the supplyvoltage calibration mode.
 15. A method for measuring a supply voltage,the method comprising: comparing the supply voltage to a referencevoltage to generate a comparison signal; generating a programmablecurrent supplied to the reference voltage; generating first controlsignals for controlling the programmable current during supply voltagemeasurement mode based on the comparison signal; selecting the supplyvoltage or a voltage from a diode as a selected signal using a firstmultiplexer; and providing the selected signal to a comparator thatgenerates the comparison signal using a second multiplexer.
 16. Anintegrated circuit comprising: a comparator to compare a referencevoltage to a supply voltage received from an internal node of theintegrated circuit to generate a comparison signal; a first currentsource to supply a first current for the reference voltage; a firstcontrol circuit to control the first current through the first currentsource based on the comparison signal; a second current source to supplya second current for the reference voltage; and a second control circuitto control the second current through the second current source, whereinthe comparator compares a desired internal supply voltage received froma reference voltage circuit to the reference voltage during a supplyvoltage calibration mode.
 17. The integrated circuit of claim 16,wherein the second control circuit adjusts the second current throughthe second current source during the supply voltage calibration modebased on the comparison signal to compensate for an offset voltage inthe comparator.
 18. The integrated circuit of claim 16, wherein thefirst control circuit generates control signals for controlling thefirst current through transistors in the first current source, andwherein the first control circuit adjusts values of the control signalsuntil the comparison signal changes state.
 19. The integrated circuit ofclaim 17 further comprising: a diode; and a multiplexer configurable toprovide a voltage across the diode to the comparator during atemperature measurement mode in response to a select signal, wherein thecomparator compares the voltage across the diode to the referencevoltage during the temperature measurement mode, and wherein themultiplexer is configurable to provide the supply voltage to thecomparator during a supply voltage measurement mode in response to theselect signal.
 20. The integrated circuit of claim 16 furthercomprising: a resistor coupled between the first current source and anode at ground.